Power converter with modular stages connected by floating terminals

ABSTRACT

An apparatus for electric power conversion includes a converter having a regulating circuit and switching network. The regulating circuit has magnetic storage elements, and switches connected to the magnetic storage elements and controllable to switch between switching configurations. The regulating circuit maintains an average DC current through a magnetic storage element. The switching network includes charge storage elements connected to switches that are controllable to switch between plural switch configurations. In one configuration, the switches forms an arrangement of charge storage elements in which at least one charge storage element is charged using the magnetic storage element through the network input or output port. In another, the switches form an arrangement of charge storage elements in which an element discharges using the magnetic storage element through one of the input port and output port of the switching network.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.16/931,768 filed Jul. 17, 2020, which is a continuation of U.S.application Ser. No. 16/444,428 filed Jun. 18, 2019, now U.S. Pat. No.10,917,007, which is a continuation of U.S. application Ser. No.15/618,481, filed Jun. 9, 2017, now U.S. Pat. No. 10,326,358, which is acontinuation of U.S. application Ser. No. 15/138,692, filed on Apr. 26,2016, now U.S. Pat. No. 9,712,051, which is a continuation of U.S.application Ser. No. 14/513,747, filed on Oct. 14, 2014, now U.S. Pat.No. 9,362,826, which is a continuation of U.S. application Ser. No.13/771,904, filed on Feb. 20, 2013, now U.S. Pat. No. 8,860,396, whichis a continuation of International Application No. PCT/US2012/036455,filed on May 4, 2012, which claims the benefit of the priority date ofU.S. Provisional Application No. 61/577,271, filed on Dec. 19, 2011;U.S. Provisional Application No. 61/548,360, filed on Oct. 18, 2011; andU.S. Provisional Application No. 61/482,838, filed on May 5, 2011. Thecontent of these applications is hereby incorporated by reference in itsentirety.

FIELD OF DISCLOSURE

This disclosure relates to power supplies, and in particular to powerconverters.

BACKGROUND

Many power converters include switches and one or more capacitors thatare used, for example, to power portable electronic devices and consumerelectronics. Switch-mode power converters regulate the output voltage orcurrent by switching energy storage elements (i.e. inductors andcapacitors) into different electrical configurations using a switchnetwork. Switched capacitor converters are switch-mode power convertersthat primarily use capacitors to transfer energy. In such converters,the number of capacitors and switches increases as the transformationratio increases. Switches in the switch network are usually activedevices that are implemented with transistors. The switch network may beintegrated on a single or on multiple monolithic semiconductorsubstrates, or formed using discrete devices.

Typical DC-DC converters perform voltage transformation and outputregulation. This is usually done in a single-stage converter such as abuck converter. However it is possible to split these two functions intotwo specialized stages, namely a transformation stage, such as aswitching network, and a separate regulation stage, such as a regulatingcircuit. The transformation stage transforms one voltage into another,while the regulation stage ensures that the voltage and/or currentoutput of the transformation stage maintains desired characteristics.

For example, referring to FIG. 1, in one converter 10, a switchingnetwork 12A is connected to a voltage source 14 at an input end thereof.An input of a regulating circuit 16A is then connected to an output ofthe switching network 12A. A load 18A is then connected to an output ofthe regulating circuit 16A. Power flows between the voltage source 14and the load 18A in the direction indicated by the arrows. Such aconverter is described in US Patent Publication 2009/0278520, filed onMay 8, 2009, the contents of which are herein incorporated by reference.

SUMMARY

In one aspect, the invention features an apparatus for electric powerconversion. Such an apparatus includes a converter having an inputterminal and an output terminal. The converter includes a regulatingcircuit having an inductance, and switching elements connected to theinductance. These switching elements are controllable to switch betweenswitching configurations. The regulating circuit maintains an average DCcurrent through the inductance. The converter also includes a switchingnetwork having an input port and an output port. This switching networkincludes charge storage elements and switching elements connected to thecharge storage elements. These switching elements are controllable toswitch between switch configurations. In one switch configuration, theswitching elements form a first arrangement of charge storage elementsin which a charge storage element is charged through one of the inputport and the output port of the switching network. In anotherconfiguration, the switching elements form a second arrangement ofcharge storage elements in which a charge storage element is dischargedthrough one of the input port and output port of the switching network.The switching network and regulating circuit also satisfy at least oneof the following configurations: (1) the regulating circuit is connectedbetween the output terminal of the converter and the switching network,the switching network being an adiabatically charged switching network;(2) the regulating circuit is connected between the output terminal ofthe converter and the switching network, wherein either the switchingnetwork is a multiphase switching network, the switching network and theregulating circuit are bidirectional, or the regulator circuit ismulti-phase; (3) the regulating circuit is connected between the inputterminal of the converter and an input port of the switching network,the switching network being an adiabatically charged switching network;(4) the regulating circuit is connected between the input terminal ofthe converter and an input port of the switching network, and either theswitching network is a multiphase switching network, the switchingnetwork and the regulating circuit are bidirectional, or the regulatorcircuit is multi-phase; (5) the switching circuit is connected betweenthe regulating circuit and an additional regulating circuit; or (6) theregulating circuit is connected between the switching network and anadditional switching network.

Embodiments of the invention include those in which the switchingnetwork includes a reconfigurable switching network and those in whichthe switching network includes a multi-phase switching network.

Other embodiments include those in which the regulating circuit includesa bidirectional regulating circuit those in which the regulating circuitincludes a multi-phase regulating circuit, those in which the regulatingcircuit is bidirectional and includes a switch-mode power converter,those in which the regulating circuit is bidirectional regulatingcircuit and includes a resonant power converter, those in which theregulating circuit is connected to an output of the switching network,and those in which the regulating circuit is connected between theoutput terminal of the converter and the switching network, theswitching network being an adiabatically charged switching network.

In other embodiments, the regulating circuit is connected between theoutput terminal of the converter and a switching network, and either theswitching network is a multi-phase switching network, the switchingnetwork and the regulating circuit are bidirectional, or the regulatorcircuit is multi-phase.

In other embodiments, the regulating circuit is connected between theinput terminal of the converter and an input port of the switchingnetwork, the switching network being an adiabatically charged switchingnetwork.

In yet other embodiments, the regulating circuit is connected betweenthe input terminal of the converter and an input port of the switchingnetwork, and either the switching network is a multi-phase switchingnetwork, the switching network and the regulating circuit arebidirectional, or the regulator circuit is multi-phase.

Among the embodiments of the invention are those in which the switchingcircuit is connected between the regulating circuit and an additionalregulating circuit, and those in which the regulating circuit isconnected between the switching network and an additional switchingnetwork.

In additional embodiments, the switching circuit is configured as an ACswitching circuit. Among these embodiments are those that also include apower-factor correction circuit connected to the AC switching circuit.Among these embodiments are those in which this power-factor correctioncircuit is connected between the AC switching circuit and the regulatingcircuit.

In another aspect, the invention features an apparatus including aconverter having an input terminal and an output terminal. The converterincludes a switching network having an input port and output port. Thisswitching network includes charge storage elements, and switchingelements connected to the charge storage elements. The switchingelements are controllable to arrange the charge storage elements into aselected configuration. In at least one configuration, the switchingelements form a first group of charge storage elements for dischargingthe charge storage elements through the output port of the switchingnetwork. In another, the switching elements form a second group ofcharge storage elements for charging the charge storage elements throughthe input port of the switching network. The converter also includes abi-directional regulating circuit connected between at least one of aninput terminal of the converter and an input port of the switchingnetwork and an output terminal of the converter and an output port ofthe switching network.

In some embodiments, the switching network includes a multi-phaseswitching network.

Also included among the embodiments are those in which the bidirectionalregulating circuit includes a buck/boost circuit and those in which thebidirectional regulating circuit includes a split-pi circuit.

In another aspect, the invention features a converter having an inputterminal and an output terminal. The converter includes a switchingnetwork having an input port and output port, charge storage elements,and switching elements connected to the charge storage elements forarranging the charge storage elements into one of a plurality ofconfigurations. In one configuration, the switching elements form afirst group of charge storage elements for discharging the chargestorage elements through the output port of the switching network. Inanother configuration, the switching elements form a second group ofcharge storage elements for charging the charge storage elements throughthe input port of the switching network. The converter further includesa regulating circuit configured to provide a stepped-up voltage andconnected between the output terminal of the converter and an outputport of the switching network.

In yet another aspect, the invention features an apparatus having aninput terminal and output terminal, and a switching network having aninput port and output port, charge storage elements, and switchingelements connected to the charge storage elements. The switchingelements are controllable for causing the switching elements to bearranged in a plurality of configurations. In one configuration, theswitching elements form a first group of charge storage elements fordischarging the charge storage elements through the output port of theswitching network. In another configuration the switching elements forma second group of charge storage elements for charging the chargestorage elements through the input port of the switching network. Theapparatus further includes a source regulating circuit connected betweenan input terminal of the converter and an input port of the switchingnetwork.

Some embodiments also include a load regulating circuit connectedbetween an output terminal of the converter and an output port of theswitching network.

In another aspect, the invention features a manufacture includingmultiple switching networks and regulating circuits having inputs andoutputs that permit modular interconnections thereof for assembly of aDC-DC converter.

In some embodiments, at least one switching network includes a switchedcapacitor network. Among these are those in which the switched capacitornetwork includes an adiabatically charged switched capacitor network.These embodiments also include those in which the adiabatically chargedswitched capacitor network includes a cascade multiplier. In some ofthese embodiments, the cascade multiplier is driven by complementaryclocked current sources.

In other embodiments, at least one regulating circuit includes a linearregulator.

Embodiments also include those in which the DC-DC converter includesseries-connected switched capacitor networks, and those in which theDC-DC converter includes multiple regulating circuits that share acommon switching network.

These and other features of the invention will be apparent from thefollowing detailed description and the accompanying figures, in which:

DESCRIPTION OF THE FIGURES

FIG. 1 shows a known DC-DC converter with separate regulating circuitand switching network;

FIG. 1A shows a bidirectional version of FIG. 1;

FIGS. 2-4 show DC-DC converters with alternate configurations ofregulating circuits and switching networks;

FIG. 5 shows a particular implementation of the power converterillustrated in FIG. 4;

FIG. 6 shows an embodiment with multiple regulating circuits;

FIG. 7 shows an RC circuit;

FIG. 8 shows a model of a switched capacitor DC-DC converter;

FIGS. 9A and 9B show a series-parallel SC converter operating in chargephase and discharge phase respectively;

FIG. 10 shows a series pumped symmetric cascade multiplier with diodes;

FIG. 11 shows a parallel pumped symmetric cascade multiplier withdiodes;

FIG. 12 shows charge pump signals;

FIG. 13 shows a two-phase symmetric series pumped cascade multiplierwith switches;

FIG. 14 shows a two-phase symmetric parallel pumped cascade multiplierwith switches;

FIG. 15 shows four different cascade multipliers along withcorresponding half-wave versions;

FIG. 16 shows output impedance of a switched-capacitor converter as afunction of frequency;

FIG. 17 shows a particular implementation of the DC-DC converterillustrated in FIG. 1A with a full-wave adiabatically charged switchingnetwork;

FIG. 18 shows the DC-DC converter illustrated in FIG. 17 during phase A;

FIG. 19 shows the DC-DC converter illustrated in FIG. 17 during phase B;

FIG. 20 shows various waveforms associated with a 4:1 adiabaticallycharged converter;

FIG. 21 shows adiabatic charging of series connected stages;

FIG. 22 shows a particular implementation of the power converterillustrated in FIG. 21;

FIG. 23 shows an AC voltage rectified using a reconfiguredswitched-capacitor stage;

FIG. 24 shows an AC-DC power converter architecture;

FIG. 25 shows a particular implementation of the AC-DC converterillustrated in FIG. 24;

FIG. 26 shows the AC-DC converter illustrated in FIG. 25 during thepositive portion of the AC cycle:

FIG. 27 shows the AC-DC converter illustrated in FIG. 25 during thenegative portion of the AC cycle;

FIG. 28 shows an AC-DC power converter architecture with power-factorcorrection;

FIGS. 29 and 30 show particular implementations of the DC-DC converterillustrated in FIG. 1;

FIGS. 31 and 32 show particular implementations of the DC-DC converterillustrated in FIG. 3;

FIGS. 33 and 34 show particular implementations of the DC-DC converterillustrated in FIG. 2; and

FIGS. 35 and 36 show particular implementations of the DC-DC converterillustrated in FIG. 4.

DETAILED DESCRIPTION

Embodiments described herein rely at least in part on the recognitionthat in a multi-stage DC-DC converter, a switching network and aregulating circuit can be made essentially modular and can be mixed andmatched in a variety of different ways. This provides a transformativeintegrated power solution (TIPS™) for the assembly of such converters.As such, the configuration shown in FIG. 1 represents only one ofmultiple ways to configure one or more switching networks 12A with oneor more regulating circuits 16A. FIG. 1A shows a bidirectional versionof FIG. 1, where power can flow either from a voltage source 14 to aload ISA or from the load ISA to the voltage source as indicated by thearrows.

There are two fundamental elements described in connection with thefollowing embodiments: switching networks and regulating circuits.Assuming series connected elements of the same type are combined, thereare a total of four basic building blocks. These are shown FIGS. 1-4.The embodiments disclosed herein include at least one of the four basicbuilding blocks shown in FIGS. 1-4.

Additional embodiments further contemplate the application ofobject-oriented programming concepts to the design of DC-DC convertersby enabling switching networks 12A and regulating circuits 16A to be“instantiated” in a variety of different ways, so long as their inputsand outputs continue to match in a way that facilitates modular assemblyof DC-DC converters having various properties.

The switching network 12A in many embodiments is instantiated as aswitching capacitor network. Among the more useful switched capacitortopologies are: Ladder, Dickson, Series-Parallel, Fibonacci, andDoubler, all of which can be adiabatically charged and configured intomulti-phase networks. A particularly useful switching capacitor networkis an adiabatically charged version of a full-wave cascade multiplier.However, diabatically charged versions can also be used.

As used herein, changing the charge on a capacitor adiabatically meanscausing an amount of charge stored in that capacitor to change bypassing the charge through a non-capacitive element. A positiveadiabatic change in charge on the capacitor is considered adiabaticcharging while a negative adiabatic change in charge on the capacitor isconsidered adiabatic discharging. Examples of non-capacitive elementsinclude inductors, magnetic elements, resistors, and combinationsthereof.

In some cases, a capacitor can be charged adiabatically for part of thetime and diabatically for the rest of the time. Such capacitors areconsidered to be adiabatically charged. Similarly, in some cases, acapacitor can be discharged adiabatically for part of the time anddiabatically for the rest of the time. Such capacitors are considered tobe adiabatically discharged.

Diabatic charging includes all charging that is not adiabatic anddiabatic discharging includes all discharging that is not adiabatic.

As used herein, an adiabatically charged switching network is aswitching network having at least one capacitor that is bothadiabatically charged and adiabatically discharged. Adiabaticallycharged switching network is a switching network that is not anadiabatically charged switching network.

The regulating circuit 16A can be instantiated as any converter with theability to regulate the output voltage. A buck converter for example, isan attractive candidate due to its high efficiency and speed. Othersuitable regulating circuits 16A include boost converters, buck/boostconverters, fly-back converters, Cuk converters, resonant converters,and linear regulators.

In one embodiment, shown in FIG. 2, a voltage source 14 provides aninput to a first switching network 12A, which is instantiated as aswitched capacitor network. The output of the first switching network12A is a lower voltage than the input voltage that is provided to aregulating circuit 16A (e.g. a buck, a boost, or a buck/boostconverter). This regulating circuit 16A provides a regulated inputvoltage to a second switching network 12B, such as another switchedcapacitor network. A high voltage output of this second switchingnetwork 12B is then applied to a load 18A.

An embodiment such as that shown in FIG. 2 can be configured to regulatethe load 18A or to regulate the source 14 depending on the direction ofenergy flow.

In another embodiment, shown in FIG. 3, a low voltage source 14 connectsto an input of a regulating circuit 16A, the output of which is providedto an input of a switching network 12A to be boosted to a higher DCvalue. The output of the switching network is then provided to a load18A.

An embodiment such as that shown in FIG. 3 can be used to regulate thesource or the load 18A depending on the direction of energy flow.

Referring now to FIG. 4, another embodiment of a converter 100 includesa first regulating circuit 300A connected to an input 102 thereof and asecond regulating circuit 300B connected to an output 104 thereof.Between the first and second regulating circuits 300A, 300B is aswitching network 200 having an input 202 and an output 204. Theswitching network includes charge storage elements 210 interconnected byswitches 212. These charge storage elements 210 are divided into firstand second groups 206, 208.

In some embodiments, the switching network 200 can be a bidirectionalswitching capacitor network such as that shown in FIG. 5. The switchingcapacitor network in FIG. features a first capacitor 20 and a secondcapacitor 22 in parallel. A first switch 24 selectively connects one ofthe first and second capacitors 20, 22 to a first regulating circuit300A, and a second switch 26 selectively connects one of the first andsecond capacitors 20, 22 to the second regulating circuit 300B. Both thefirst and second switches 24, 26 can be operated at high frequency, thusfacilitating the adiabatic charging and discharging of the first andsecond capacitors 20, 22.

The particular embodiment shown in FIG. 5 has a two-phase switchingnetwork 200. However, other types of switching networks can be usedinstead.

In yet another embodiment, shown in FIG. 6, multiple regulating circuits16A, 16B, 16C are provided at an output of a first switching network 12Afor driving multiple loads 18A-18C. For one of the loads 18C, a secondswitching network 12B is provided between the load 18C and thecorresponding regulating circuit 16C thus creating a pathway similar tothat shown in FIG. 2. FIG. 6 thus provides an example of how the modularconstruction of regulating circuits and switching networks facilitatesthe ability to mix and match components to provide flexibility in DC-DCconverter construction.

A switched capacitor (SC) DC-DC power converter includes a network ofswitches and capacitors. By cycling the network through differenttopological states using these switches, one can transfer energy from aninput to an output of the SC network. Some converters, known as “chargepumps,” can be used to produce high voltages in FLASH and otherreprogrammable memories.

FIG. 7 shows a capacitor C initially charged to some value V_(C)(0). Att=0 the switch S is closed. At that instant, a brief surge of currentflows as the capacitor C charges to its final value of V_(m). The rateof charging can be described by a time constant τ=RC, which indicatesthe time it takes the voltage to either rise or fall to within 1/e ofits final value. The exact capacitor voltage v_(c)(t) and currenti_(c)(t) are given by the following equations:

$\begin{matrix}{{{v_{c}(t)} = {{v_{c}(0)} + {\left\lbrack {V_{in} - {v_{c}(0)}} \right\rbrack\left( {1 - e^{{- t}/{RC}}} \right)}}},{and}} & (1.1) \\{{i_{c}(t)} = {{C\frac{{dv}_{c}}{dt}} = {\frac{V_{in} - {v_{c}(0)}}{R}{e^{{- t}/{RC}}.}}}} & (1.2)\end{matrix}$

The energy loss incurred while charging the capacitor can be found bycalculating the energy dissipated in resistor R, which is

$\begin{matrix}{{{E_{loss}(t)} = {\int_{t = 0}^{\infty}{{i_{R}(t)} \times {v_{R}(t)}}}}{{dt} = {\int_{t = 0}^{\infty}{\left\lbrack {i_{c}(t)} \right\rbrack^{2}R\mspace{14mu}{{dt}.}}}}} & (1.3)\end{matrix}$

The equation can be further simplified by substituting the expressionfor i_(c)(t) from equation (1.2) into equation (1.3). Evaluating theintegral then yields

${E_{loss}(t)} = {{\frac{1}{2}\left\lbrack {V_{in} - {v_{c}(0)}} \right\rbrack}^{2}{{C\left\lbrack {1 - e^{{- 2}\;{t/{RC}}}} \right\rbrack}.}}$

If the transients are allowed to settle (i.e. t→∞), the total energyloss incurred in charging the capacitor is independent of its resistanceR. In that case, the amount of energy loss is equal to

${E_{loss}(\infty)} = {\frac{1}{2}C\;\Delta\; v_{c}^{2}}$

A switched capacitor converter can be modeled as an ideal transformer,as shown in FIG. 8, with a finite output resistance R_(o) that accountsfor the power loss incurred in charging or discharging of the energytransfer capacitors, as shown in FIG. 8. This loss is typicallydissipated in the ON resistance of the MOSFETs and equivalent seriesresistance of the capacitors.

The output voltage of the switched-capacitor converter is given by

$V_{o} = {{V_{in}\frac{N_{2}}{N_{1}}} - {I_{o}R_{o}}}$

There are two limiting cases where the operation of the switchedcapacitor converters can be simplified and R_(o) easily found. These arereferred to as the “slow-switching limit” and the “fast-switchinglimit.”

In the fast-switching limit (τ>>T_(sw)), the charging and dischargingcurrents are approximately constant, resulting in a triangular AC rippleon the capacitors. Hence, R_(o) is sensitive to the series resistance ofthe MOSFETs and capacitors, but is not a function of the operatingfrequency. In this case, the output resistance of the converteroperating in the fast-switching limit is a function of parasiticresistance.

In the slow-switching limit, the switching period T_(sw) is much longerthan the RC time constant τ of the energy transfer capacitors. Underthis condition, systemic energy loss irrespective of the resistance ofthe capacitors and switches. This systemic energy loss arises in partbecause the root mean square (RMS) of the charging and dischargingcurrent is a function of the RC time constant. If the effectiveresistance R_(eff) of the charging path is reduced (i.e. reduced RC),the RMS current increases and it so happens that the total chargingenergy loss (E_(loss)=I_(RMS) ²R_(eff)=1/2C×ΔV_(C2)) is independent ofR_(eff). One solution to minimize this energy loss is to increase thesize of the pump capacitors in the switched capacitor network.

It is desirable for a switching capacitor network to have a commonground, large transformation ratio, low switch stress, low DC capacitorvoltage, and low output resistance. Among the more useful topologiesare: Ladder, Dickson, Series-Parallel, Fibonacci, and Doubler.

One useful converter is a series-parallel switched capacitor converter.FIGS. 9A and 9B show a 2:1 series-parallel switched capacitor converteroperating in charge phase and in discharge phase, respectively. Duringthe charge phase, the capacitors are in series. In the discharge phase,the capacitors are in parallel. In its charge phase, capacitor voltagesv_(C1) and v_(C2) add up to V₁ while in its discharge phase, v_(C1) andv_(C2) equal V₂, which means that V₂=V₁/2.

Other useful topologies are cascade multiplier topologies, as shown inFIGS. 10 and 11. In both charge pumps, the source is located at V₁ andthe load is located at V₂. In these types of charge pumps, packets ofcharge are pumped along a diode chain as the coupling capacitors aresuccessively charged and discharged. As shown in FIG. 12, clock signalsν_(clk) and ν_(clk) with amplitude v_(pump) are 180 degrees out ofphase. The coupling capacitors can either be pumped in series orparallel.

It takes n clock cycles for the initial charge to reach the output. Thecharge on the final pump capacitor is n times larger than the charge onthe initial pump capacitor and thus the output voltage V₂ for theconverters is V₁+(n−1)×v_(pump) in both pumping configurations.

Although the foregoing topologies are suitable for stepping up voltage,they can also be used to step down voltage by switching the location ofthe source and the load. In such cases, the diodes can be replaced withcontrolled switches such as MOSFETs and BJTs.

The foregoing cascade multipliers are half-wave multipliers in whichcharge is transferred during one phase of the of the clock signal. Thiscauses a discontinuous input current. Both of these cascade multiplierscan be converted into full-wave multipliers by connecting two half-wavemultipliers in parallel and running the half-wave multipliers degreesout of phase. FIG. 13 shows a full-wave symmetric series pumped cascademultiplier version while FIG. 14 shows a full-wave symmetric parallelpumped cascade multiplier version. Unlike the diodes in thehalf-multiplier, the switches in FIG. 13 and FIG. 14 are bidirectional.As a result, in both of these cascade multipliers, power can flow eitherfrom the source to the load or from the load to the source. Asymmetricmultipliers can also be converted into full-wave multipliers

FIG. 15 shows four different step-up versions of full-wave symmetricmultipliers along with their corresponding half-wave versions.Furthermore, it is possible to combine N phases in parallel and run them180 degrees/N out of phase to reduce output voltage ripple and increaseoutput power handling capability.

The basic building blocks in the modular architecture shown FIGS. 1-4can either be connected as independent entities or coupled entities. Inthe situation where the switching networks and regulating circuits aretightly coupled, it is possible to prevent and/or reduce the systemicenergy loss mechanism of the switching networks through adiabaticcharging. This generally includes using a regulating circuit to controlthe charging and discharging of the capacitors in the switching network.Furthermore, the output voltage of the regulating circuit and thus thetotal converter can be regulated in response to external stimuli. Oneapproach to regulating the output voltage is by controlling the averageDC current in the magnetic storage element.

A desirable feature of a regulating circuit is to limit the root meansquare (RMS) current through the capacitors in the switching network. Todo that, the regulating circuit uses either resistive or magneticstorage elements. Unfortunately, resistive elements would consume powerso their use is less desirable. Therefore, embodiments described hereinrely on a combination of switches and a magnetic storage element in theregulating circuit. The regulating circuit limits the RMS current byforcing the capacitor current through a magnetic storage element in aregulating circuit that has an average DC current. The switches in theregulating circuit are operated so as to maintain an average DC currentthrough the magnetic storage element.

The regulating circuit may limit both the RMS charging current and theRMS discharging current of at least one capacitor in the switchingnetwork. A single regulating circuit may limit the current in or out ofswitching network by sinking and/or sourcing current. Therefore, thereare four fundamental configurations, which are shown in FIGS. 1-4.Assuming power flows from source to load then, in FIG. 1, regulatingcircuit 16A may sink both the charging and discharging current ofswitching network 12A. In FIG. 3, regulating circuit 16A may source boththe charging and discharging current of switching network 12A. In FIG.4, regulating circuit 300A may source the charging current of switchingnetwork 200 and regulating circuit 300B may sink the discharging currentof the same switching network 200 and vice-versa. In FIG. 2, regulatingcircuit 16A may source both the charging and discharging current ofswitching network 12B while also sinking both the charging anddischarging current of switching network 12A. Furthermore, if both theswitching networks and regulating circuits allow power to flow in bothdirections then bidirectional power flow is possible (source to load andload to source).

One embodiment relies on at least partially adiabatically chargingfull-wave cascade multipliers. Cascade multipliers are a preferredswitching network because of their superior fast-switching limitimpedance, ease of scaling up in voltage, and low switch stress.

In cascade multipliers, the coupling capacitors are typically pumpedwith a clocked voltage source ν_(clk) & ν_(clk) . However, if thecoupling capacitors are pumped with a clocked current source i_(clk) &i_(clk) instead, then the RMS charging and discharging current in thecoupling capacitor may be limited. In this case, the capacitors are atleast partially charged adiabatically thus lowering, if not eliminating,the 1/2C×ΔV_(c) ² loss that is associated with a switched capacitorconverter when operated in the slow-switching limit. This has the effectof lowering the output impedance to the fast-switching limit impedance.As shown by the black dotted line in FIG. 16, which depicts adiabaticoperation under full adiabatic charging, the output impedance would nolonger be a function of switching frequency.

With all else being equal, an adiabatically charged switched-capacitorconverter can operate at a much lower switching frequency than aconventionally charged switched capacitor converter, but at higherefficiency. Conversely, an adiabatically charged switched-capacitorconverter can operate at the same frequency and with the same efficiencyas a conventionally charged switched-capacitor converter, but with muchsmaller coupling capacitors, for example between four and ten timessmaller.

FIG. 17 shows a step-down converter consistent with the architectureshown in FIG. 1A. However, in this embodiment, the switching network 12Ais adiabatically charged using the regulating circuit 16A. The clockedcurrent sources i_(clk) & i_(clk) are emulated by four switches andregulating circuit 16A. The output capacitor C_(O) has also been removedso as to allow V_(X) to swing. In this example, the regulating circuit16A is a boost converter that behaves as constant source with a small ACripple. Any power converter that has a non-capacitive input impedancewould have allowed adiabatic operation. Although switch-mode powerconverters are attractive candidates due to their high efficiency,linear regulators are also practical.

In operation, closing switches labeled 1 charges capacitors C₄, C₅, andC₆ while discharging capacitors C₁, C₂ and C₃. Similarly, closingswitches 2 has the complementary effect. The first topological state(phase A) is shown in FIG. 18, where all switches labeled 1 are closedand all switches labeled 2 are opened. Similarly, the second topologicalstate (phase B) is shown in FIG. 19, where all switches labeled 2 areclosed and all switches labeled 1 are opened. In this embodiment, theregulating circuit 16A limits the RMS charge and discharging current ofeach capacitor. For example, capacitor C₃ is discharged through thefilter inductor in regulating circuit 16A during phase A, whilecapacitor C₃ is charged through the filter inductor in regulatingcircuit 16A during phase B, clearly demonstrating the adiabatic concept.Furthermore, all of the active components are implemented with switchesso the converter can process power in both directions.

A few representative node voltages and currents are shown in FIG. 20.There is a slight amount of distortion on the rising and falling edgesof the two illustrated currents (I_(P1) and I_(P2)), but for the mostpart, the currents resemble two clocks 180 degrees out of phase. Ingeneral, adiabatic charging occurs in cascade multipliers if at leastone end of a switch stack is not loaded with capacitance, as is the casein this embodiment, where the V_(x) node is loaded down by theregulating circuit 16A.

The modular architecture with the basic building blocks shown in FIGS.1-4 may be expanded to cover a wider range of applications, such ashigh-voltage DC, AC-DC, buck-boost, and multiple output voltages. Eachof these applications includes separating the transformation andregulation functions. Extension of the architecture can also incorporateadiabatically charged switched capacitors converters.

In many switched-capacitor converters, the number of capacitors andswitches increases linearly with the transformation ratio. Thus, a largenumber of capacitors and switches are required if the transformationratio is large. Alternatively, a large transformation ratio can beachieved by connecting numerous low gain stages in series as depicted inFIG. 21. The transformation ratio of the total switch capacitor stack(V_(m)/V_(x)) is as follows:

$\begin{matrix}{\frac{V_{in}}{V_{x}} = {N_{1} \times N_{2}\mspace{14mu}\ldots\mspace{14mu} N_{n}}} & (2.1)\end{matrix}$

The main disadvantage of the series stacked configuration is that thevoltage stresses on the front stages are much higher than those of therear stages. This will normally require stages with different voltageratings and sizes.

Adiabatic charging of a preceding series-connected switching networkonly occurs if the following switching network controls the charging anddischarging current of the preceding stage. Thus, it is preferable touse full-wave switched-capacitor converters in the front stages or touse switched-capacitor stages such as the single-phase series-parallelswitched-capacitor converters with magnetic based filters.

FIG. 22 shows a converter with two series-connected switching networksconsistent with the architecture shown in FIG. 21. Both switchingnetworks 12A and 12D are two-phase cascade multipliers. In operation,switches labeled 1 and 2 are always in complementary states and switcheslabeled 7 and 8 are always in complementary states. Thus, in a firstswitched-state, all switches labeled “1” are open and all switcheslabeled “2” are closed. In a second switched-state, all switches labeled“1” are closed and all switches labeled “2” are opened. In thisembodiment, closing switches 1 charges the capacitors C₁, C₂, C₃, whiledischarging the capacitors C₄, C₅, C₆ and closing switches 2 has thecomplementary effect. Also, closing switches 7 charges capacitors C₇,C₈, C₉, while discharging capacitors C₁₀, C₁₁, C₁₂ and closing switches8 has the complementary effect.

The power converter provides a total step-down of 32:1, assuming theregulating circuit 16A is a buck converter with a nominal step-downratio of 2:1. Furthermore, if the input voltage is 32 V and the outputvoltage is 1 V, then the switches in the first switching network 12Awill need to block 8 volts while the switches in the second switchingnetwork 12D will need to block 2 volts.

The modular architecture with the basic building blocks shown in FIGS.1-4 may be configured to handle an AC input voltage as well. One of themain attributes of switched capacitor converters is their ability tooperate efficiency over a large input range by reconfiguring theswitched-capacitor network. If the AC wall voltage (i.e. 60 Hz &V_(RMS)) can be thought of as a slow moving DC voltage, then thefront-end switched-capacitor stage should be able to unfold thetime-varying input voltage into a relatively stable DC voltage.

A diagram of a 120 V_(RMS) AC waveform over a single 60 Hz cycleoverlaid with the unfolded DC voltage is shown in FIG. 23. The ACswitching network has different configurations (1/3, 1/2, 1/1) at itsdisposal along with an inverting stage. It was also designed to keep theDC voltage under 60 V. Once the AC voltage is unfolded, it is the job ofthe regulating circuit 16A, shown in FIG. 24, to produce a final outputvoltage. It may also be necessary to place another switching network 16Abetween the AC switching network 13A and regulating circuit 16A tofurther condition the voltage. If this is the case, then the caveats forseries-connected stages hold true since the AC switching network 13A isa special purpose switching network 12A.

FIG. 25 shows the AC-DC converter corresponding to the architectureshown in FIG. 24. In this embodiment, the AC switching network 13A is asynchronous AC bridge followed by a reconfigurable two-phase step-downcascade multiplier with three distinct conversion ratios (1/3, 1/2, 1/1)while the regulating circuit 16A is a synchronous buck converter. Inoperation, switches labeled 7 and 8 are always in complementary states.During the positive portion of the AC cycle (0 to π radians) allswitches 7 are closed while all switches labeled 8 are opened as shownin FIG. 26. Similarly, during the negative portion of the AC cycle (π to2π radians) all switches labeled 8 are closed while all switches labeled7 are opened as shown in FIG. 27.

In addition to the inverting function provided by switches 7 and 8, theswitches labeled 1A-1E and switches labeled 2A-2E may be selectivelyopened and closed as shown in Table 1 to provide three distinctconversion ratios of: 1/3, 1/2 and 1.

TABLE 1 V₂/V₁ 1A 1B 1C 1D 1E 1/3 CLK CLK CLK CLK CLK 1/2 CLKB CLK CLKCLK CLK 1/1 ON ON ON OFF OFF V₂/V₁ 2A 2B 2C 2D 2E 1/3 CLKB CLKB CLKBCLKB CLKB 1/2 CLK CLKB CLKB CLKB CLKB 1/1 ON ON ON OFF OFF

The AC switching network 13A is provided with a digital clock signalCLK. A second signal CLKB is also generated, which may simply be thecomplement of CLK (i.e. is high when CLK is low and low when CLK ishigh), or which may be generated as a non-overlapping complement as iswell known in the art. With a switching pattern set in accordance withthe first row of Table 1, the AC switching network 13A provides astep-down ratio of one-third (1/3). With a switching pattern set inaccordance with the second row of Table 1, the AC switching network 13Aprovides a step-down ratio of one-half (1/2). With a switching patternset in accordance with the first row of Table 1, the AC switchingnetwork 13A provides a step-down ratio of one.

Most power supplies attached to the wall meet some power factorspecification. Power factor is a dimensionless number between 0 and 1that defines a ratio of the real power flowing to apparent power. Acommon way to control the harmonic current and thus boost the powerfactor is by using an active power factor corrector, as shown in FIG.28. The power-factor correction circuit 17A causes the input current tobe in phase with the line voltage, thus causing reactive powerconsumption to be zero.

FIGS. 29-36 show specific implementations of power converters thatconform to the architectural diagrams shown in FIGS. 1-4. In eachimplementation a regulating circuit or multiple regulating circuits maylimit both the RMS charging current and the RMS discharging current ofat least one capacitor in each switching network so all of theseswitching networks are adiabatically charged switching networks.However, if decoupling capacitors 9A or 9B are present, then the abilityof the regulating circuit to limit the RMS charging and dischargingcurrent may be diminished. Capacitors 9A and 9B are optional and to keepthe output voltage fairly constant capacitor C_(O) is used. Furthermore,for simplicity, the switching network in each implementation has asingle conversion ratio. However, reconfigurable switching networks thatprovide power conversion at multiple distinct conversion ratios may beused instead.

In operation, switches labeled 1 and 2 are always in complementarystates. Thus, in a first switched-state, all switches labeled “1” areopen and all switches labeled “2” are closed. In a secondswitched-state, all switches labeled “1” are closed and all switcheslabeled “2” are opened. Similarly, switches labeled “3” are “4” are incomplementary states, switches labeled “5” are “6” are in complementarystates, and switches labeled “7” are “8” are in complementary states.Typically, the regulating circuits operate at higher switchingfrequencies than the switching networks. However, there is norequirement on the switching frequencies between and amongst theswitching networks and regulating circuits.

FIG. 29 shows a step-up converter corresponding to the architectureshown in FIG. 1. In this embodiment, the switching network 12A is atwo-phase step-up cascade multiplier with a conversion ratio of 1:3while the regulating circuit 16A is two-phase boost converter. Inoperation, closing switches labeled 1 and opening switches labeled 2charges capacitors C₃ and C₄ while discharging capacitors C₁ and C₂.Conversely, opening switches labeled 1 and closing switches labeled 2charges capacitors C₁, and C₂ while discharging capacitors C₃ and C₄.

FIG. 30 shows bidirectional step-down converter corresponding to thearchitecture shown in FIG. 1A. In this embodiment, the switching network12A is a two-phase step-down cascade multiplier with a conversion ratioof 4:1 while the regulating circuit 16A is synchronous buck converter.In operation, closing switches labeled 1 and opening switches labeled 2charges capacitors C₁, C₂, and C₃ while discharging capacitors C₄, C₅,and C₆. Conversely, opening switches labeled 1 and closing switcheslabeled 2 charges capacitors C₄, C₅, and C₆ while discharging capacitorsC₁, C₂, and C₃. All of the active components are implemented withswitches so the converter can process power in both directions.

FIG. 31 shows a step-up converter consistent with the architecture shownin FIG. 3. In this embodiment, the regulating circuit 16A is boostconverter while the switching network 12A is a two-phase step-upseries-parallel SC converter with a conversion ratio of 2:1. Inoperation, closing switches 1 charges capacitor C₂ while dischargingcapacitor C₁. Closing switches 2 has the complementary effect.

FIG. 32 shows a bidirectional up-down converter consistent with thearchitecture shown in FIG. 3. In this embodiment, the regulating circuit16A is synchronous four switch buck-boost converter while the switchingnetwork 12A is a two-phase step-up cascade multiplier with a conversionratio of 4:1. In operation, closing switches labeled 1 chargescapacitors C₄, C₅, and C₆ while discharging capacitors C₁, C₂, and C₃.Closing switches 2 has the complementary effect. All of the activecomponents are implemented with switches so the converter can processpower in both directions.

FIG. 33 shows an inverting up-down converter consistent with thearchitecture shown in FIG. 2. In this embodiment, the switching network12A is a step-up series-parallel SC converter with a conversion ratio of2:1, the regulating circuit 16A is a buck/boost converter and theswitching network 12B is a step-up series-parallel SC converter with aconversion ratio of 2:1. In operation, closing switches 1 chargescapacitor C₁ while closing switches 2 discharges capacitor C₁.Similarly, closing switches discharges capacitor C₂ while closingswitches 8 charges capacitor C₂.

FIG. 34 shows a bidirectional inverting up-down converter consistentwith the architecture shown in FIG. 2. In this embodiment, the switchingnetwork 12A is a two-phase step-up series-parallel SC converter with aconversion ratio of 2:1, the regulating circuit 16A is a synchronousbuck/boost converter and the switching network 12B is a two-phasestep-up series-parallel SC converter with a conversion ratio of 2:1. Inoperation, closing switches 1 charges capacitor C₁ while dischargingcapacitor C₂. Closing switches 2 has the complementary effect.Similarly, closing switches 7 charges capacitor C₄ while dischargingcapacitor C₃. Closing switches 2 has the complementary effect. All ofthe active components are implemented with switches so the converter canprocess power in both directions.

FIG. 35 shows a step-down converter consistent with the block diagramshown in FIG. 4. In this embodiment, the regulating 300A is a boostconverter, the switching network 200 is a two-phase step-upseries-parallel SC converter with a conversion ratio of 2:1 and theregulating circuit 300B is a boost converter. In operation, closingswitches labeled 1 charges capacitors C₁ and C₂ while simultaneouslydischarging capacitors C₃ and C₄. Closing switches 2 has thecomplementary effect.

FIG. 36 shows a bidirectional up-down converter consistent with theblock diagram shown in FIG. 4. In this embodiment, the regulating 300Ais a synchronous boost converter, the switching network 200 is atwo-phase fractional step-down series-parallel SC converter with aconversion ratio of 3:2 and the regulating circuit 300B is a synchronousbuck converter. In operation, closing switches 1 charges capacitors C₃and C₄ while simultaneously discharging capacitors C₁ and C₂. Closingswitches 2 has the complementary effect. All of the active componentsare implemented with switches so the converter can process power in bothdirections.

It should be understood that the topology of the regulating circuit canbe any type of power converter with the ability to regulate the outputvoltage, including, but without limitation, synchronous buck,three-level synchronous buck, SEPIC, soft switched or resonantconverters. Similarly, the switching networks can be realized with avariety of switched-capacitor topologies, depending on desired voltagetransformation and permitted switch voltage.

Having described one or more preferred embodiments, it will be apparentto those of ordinary skill in the art that other embodimentsincorporating these circuits, techniques and concepts may be used.Accordingly, it is submitted that the scope of the patent should not belimited to the described embodiments, but rather, should be limited onlyby the spirit and scope of the appended claims.

The invention claimed is:
 1. An apparatus, comprising: a switchedcapacitor power converter having an input port and an output port, theswitched capacitor power converter comprising: one or more drivers; aplurality of switches to be interconnected with a plurality of passivedevices and controllable to switch between at least a first switchconfiguration and a second switch configuration to respectivelycorrespond to at least a first conduction path and a second conductionpath; a controller operable to alternate switching the plurality ofswitches in accordance with one or more switching frequencies totransfer energy from the input port to the output port of switchedcapacitor power converter; and a modulator to generate a switchingfrequency of the one or more switching frequencies to facilitate softswitching of at least some of the plurality of switches, wherein thefirst conduction path via the first switch configuration is tofacilitate at least a positive change in charge on at least one passivedevice of the plurality of passive devices and the second conductionpath via the second switch configuration is to facilitate at least anegative change in charge on the at least one passive device of theplurality of passive devices.
 2. The apparatus of claim 1, wherein themodulator to generate the switching frequency of the one or moreswitching frequencies to facilitate the soft switching of the at leastsome of the plurality of switches to comprise a modulator to generatethe switching frequency of the one or more switching frequencies tofacilitate zero current switching (ZCS) of the at least some of theplurality of switches.
 3. The apparatus of claim 1, and furthercomprising a memory.
 4. The apparatus of claim 3, wherein the memorycomprises a programmable memory.
 5. The apparatus of claim 4, whereinthe programmable memory to be used, at least in part, to facilitateoperation of the switched capacitor power converter.
 6. The apparatus ofclaim 3, wherein the memory comprises a reprogrammable memory.
 7. Theapparatus of claim 6, wherein the reprogrammable memory comprises aFLASH memory.
 8. The apparatus of claim 6, wherein the switchedcapacitor power converter to be used, at least in part, to produce avoltage in the reprogrammable memory.
 9. The apparatus of claim 1,wherein the energy is to be provided to the input port during operationof the switched capacitor power converter by a source of electricalenergy.
 10. The apparatus of claim 1, wherein the source of electricalenergy to comprise at least one of the following: a power converter or abattery.
 11. The apparatus of claim 1, wherein the plurality of passivedevices comprises at least one of the following: one or more capacitors;one or more inductors; or any combination thereof.
 12. A power convertercomprising: a first port; a second port; at least one inductor insertedinto a power path between the first port and the second port, the atleast one inductor having respective first and second nodes to couplethe at least one inductor to the power path; and a switching network toinclude a plurality of switches, one or more switches of the pluralityof switches to interconnect one or more capacitors with the at least oneinductor via one or more interconnections during operation of the powerconverter, at least one interconnection of the one or moreinterconnections to form a switched-capacitor network to switch betweena first switch configuration and a second switch configuration, whereinthe switched-capacitor network to be controllable to switch between thefirst and the second switch configuration at one or more switchingfrequencies so as to transfer energy via the power path between thefirst port and the second port, wherein the at least one inductor tofacilitate a change in charge on at least some of the one or morecapacitors at a charge rate during operation of the power converter, thecharge rate to be determined, at least in part, by the at least oneinductor, and wherein the one or more switching frequencies to include afrequency to facilitate soft switching of at least some switches of theplurality of switches.
 13. The power converter of claim 12, wherein thefrequency to facilitate the soft switching of the at least some switchesof the plurality of switches to comprise a frequency to facilitate zerocurrent switching (ZCS) of the at least some switches of the pluralityof switches.
 14. The power converter of claim 12, wherein the at leastone inductor comprises a magnetic core.
 15. The power converter of claim14, wherein the at least one inductor to include windings supported bythe magnetic core, the windings to be disposed between the respectivefirst and second nodes of the at least one inductor.
 16. The powerconverter of claim 12, wherein the switched-capacitor network to becontrollable via a controller.
 17. The power converter of claim 16,wherein the controller is to implement at least one of the following: avariable-frequency control or control based, at least in part, oncurrent.
 18. The power converter of claim 17, wherein thevariable-frequency control is to be implemented, at least in part, viaone or more pulse-width modulation (PWM) signals.
 19. The powerconverter of claim 16, wherein the controller to implement a deadtimeinterval.
 20. The power converter of claim 19, wherein the controller toimplement the deadtime interval so as to prevent the plurality ofswitches from conducting simultaneously.
 21. The power converter ofclaim 12, wherein the at least one inductor to facilitate the change incharge on the at least some of the one or more capacitors at a dischargerate during operation of the power converter, the discharge rate to bedetermined, at least in part, by the at least one inductor.
 22. Thepower converter of claim 21, wherein the charge rate and the dischargerate are substantially the same.
 23. The power converter of claim 12,wherein the power converter is to provide power to one or more loads,directly or indirectly.
 24. The power converter of claim 23, wherein thepower is to be provided via a buck converter.
 25. The power converter ofclaim 24, wherein the buck converter comprises a multi-phase buckconverter.
 26. The power converter of claim 23, wherein the second portis to provide a voltage to be regulated via a buck converter.
 27. Aresonant power converter, comprising: one or more drivers; a controllerto implement a deadtime interval; and a switching network to include aplurality of switches controllable to switch between at least a firstand a second switch configurations so as to selectively interconnect aplurality of capacitors with a plurality of inductors, wherein theswitching network comprises a first group of switches of the pluralityof switches to interconnect first capacitors of the plurality ofcapacitors with a first inductor of the plurality of inductors to form afirst switched-capacitor network during operation of the resonant powerconverter, wherein the switching network further comprises a secondgroup of switches of the plurality of switches to interconnect secondcapacitors of the plurality of capacitors with a second inductor of theplurality of inductors to form a second switched-capacitor networkduring operation of the resonant power converter, wherein the firstswitched-capacitor network to be formed via the first switchconfiguration to correspond to a first phase and the secondswitched-capacitor network to be formed via the second switchconfiguration to correspond to a second phase, wherein the deadtimeinterval to be implemented, at least in part, to facilitate softswitching of at least some switches of the first group of switches orthe second group of switches of the resonant power converter.
 28. Theresonant power converter of claim 27, wherein the first phase via thefirst switch configuration is to facilitate at least a positive changein charge on at least one capacitor of the first capacitors and thesecond phase via the second switch configuration is to facilitate atleast a negative change in charge on the at least one capacitor of thefirst capacitors.
 29. The resonant power converter of claim 27, whereinthe deadtime to be implemented, at least in part, to deactivate theplurality of switches to prevent the plurality of switches fromconducting simultaneously.
 30. The resonant power converter of claim 27,wherein the deadtime to be implemented, at least in part, between thefirst phase and the second phase.
 31. The resonant power converter ofclaim 27, wherein the resonant power converter is to implement aparticular conversion ratio.
 32. The resonant power converter of claim31, wherein the particular conversion ratio to be determined based, atleast in part, on a number of capacitors of the plurality of capacitorsto be included in the resonant power converter.
 33. The resonant powerconverter of claim 31, wherein the particular conversion ratio comprisesat least one of the following conversion ratios: a conversion ratio of4:1; a conversion ratio of 3:1; a conversion ratio of 2:1; or aconversion ratio of 1:1.
 34. The resonant power converter of claim 27,wherein the one or more drivers to drive the plurality of switches toimplement at least one of the following: the first switch configuration;the second switch configuration; or any combination thereof.
 35. Theresonant power converter of claim 27, wherein the controller to generateone or more control signals for the resonant power converter.
 36. Theresonant power converter of claim 35, wherein the one or more controlsignals to control at least one of the following: the first group ofswitches; the second group of switches; or any combination thereof. 37.The resonant power converter of claim 27, wherein the controller is toimplement a variable-frequency control.
 38. The resonant power converterof claim 37, wherein the variable-frequency control is to beimplemented, at least in part, via a pulse-width modulator (PWM). 39.The resonant power converter of claim 27, wherein the first inductor orthe second inductor comprises a magnetic core.
 40. A power convertercomprising: a switched capacitor arrangement to include a plurality ofcapacitors to be alternately connected to a first group of switches orto a second group of switches to respectively implement a first switchconfiguration and a second switch configuration via one or moreswitching frequencies to respectively correspond to at least a firstphase and a second phase so as to transfer energy from an input port toan output port of the power converter; a switched magnetic arrangementto include at least one inductor to be arranged in a configuration withthe switched capacitor arrangement; and a controller to implement adeadtime interval, wherein the first phase via the first switchconfiguration is to facilitate a positive change in charge on at leastone capacitor of the plurality of capacitors during operation of thepower converter and the second phase via the second switch configurationis to facilitate a negative change in charge on the at least onecapacitor of the plurality of capacitors during operation of the powerconverter, and wherein the one or more switching frequencies to includea frequency to facilitate zero current switching (ZCS) of at least someswitches of the first or the second group of switches.
 41. The powerconverter of claim 40, wherein the positive change in charge comprisesan adiabatic charge and wherein the negative change in charge comprisesan adiabatic discharge.
 42. The power converter of claim 40, wherein thecontroller to is implement control based, at least in part, on current.43. The power converter of claim 40, wherein the power converter is toprovide power to multiple loads, directly or indirectly.
 44. The powerconverter of claim 43, wherein the power is to be provided via a buckconverter.
 45. The power converter of claim 40, wherein the switchedcapacitor arrangement comprises a cascade multiplier.
 46. The powerconverter of claim 45, wherein the cascade multiplier comprises aDickson charge pump.
 47. The power converter of claim 40, wherein theoutput port of the power converter is to provide a voltage to beregulated via a buck converter.